RELATED APPLICATIONS 0001 This application claims priority from U.S. Provisional Patent Application Serial No. 60/269,068, entitled High Impedance Current Mode Voltage Scalable Driver filed Feb. 15, 2001.

FIELD OF THE INVENTION 0002 The present invention relates to the field of signal conversion and, more particularly, to circuits to scale signal voltage.

BACKGROUND OF THE RELATED ART 0003 In a number of situations, a signal voltage will need to be reduced in order to couple the signal to the next circuit. Although a voltage divider network or circuit can provide a requisite step down in the voltage level of the signal, in certain applications such voltage divider reduction may not provide the adequate performance needed for the driven circuit. When the driven circuit operates at a substantially lower voltage then the driving circuit, a simple voltage divider reduction could introduce significant jitter and skew at the input of the driven circuit. For example, when the signal is a clocking signal and the driven circuit is a processor operating at a sufficiently fast speed, performance problems could be encountered if a reduction of the clocking signal is needed at the input of the processor. State of the art processors of today operate at input clock frequencies of 100 MHz or higher and these clock frequencies are multiplied within the processor chip itself. At these higher frequencies of operation, the processors may operate near or below 1.0 volt level. This is especially true of processors utilized for mobile applications where lower supply voltage for the processor core is imperative in order to conserve battery life. 0004 Although the processor technology has developed to improve the performance of the processors, clock generators have not improved upon the technology to produce lower voltage clocking circuitry. Part of the reason stems from the fact that lower voltage circuits typically are more expensive to manufacture than circuitry utilizing higher supply voltages. Accordingly, many clock vendors continue to produce clock generator chips operating at the supply voltages around 3.3 volts. In order to utilize a 3.3 volt clocking signal to drive a processor operating at a supply voltage of around 1.0 volt or below, the clock signal will need to be reduced to a fraction of its output level in order to drive the processor. Since rail-to-rail transition is much smaller for the reduced voltage signal, jitter at the input is more noticeable during the transition. Although it is possible to increase the slew rate by implementing a large device, a significant increase in the slew rate will most likely introduce undesirable electromagnetic interference (EMI) and in some instances this EMI level is beyond standards permitted for the computing devices at low voltage, it is difficult to increase the slew rate. 0005 As a further problem, some form of over-voltage protection is typically desirable in order to prevent an accidental increase in the input voltage which could damage the processor. Additionally, if the supply operating voltage of the processor drops below a volt (for example, to 0.9 volts) the processor supply voltage is approaching the threshold voltage of the clocking circuit, so that adequate signal transition may be impaired due to the closeness of the supply voltage of the processor to the threshold turn on voltage of the circuitry of the clock generator. Accordingly, for various reasons noted, a solution is needed, especially for lower supply voltage devices, such as the example processor described above.

BRIEF DESCRIPTION OF THE DRAWINGS 0006 FIG. 1 is a block schematic diagram showing one embodiment of a high impedance current mode scalable driver of the present invention. 0007 FIG. 2 is a circuit schematic diagram illustrating one embodiment of the scalable driver of the present invention. 0008 FIG. 3 is a more detailed circuit schematic diagram of an embodiment of the circuit of FIG. 2. 0009 FIG. 4 illustrates one signal waveform diagram at the output of the scalable driver of FIG. 2. 0010 FIG. 5 illustrates another waveform diagram at the output of the scalable driver of FIG. 2. 0011 FIG. 6 is a graph illustrating the current Iout versus Vout for a range of Vcc in which an effective range of operation is defined as that portion having a substantially constant current. 0012 FIG. 7 is a graph illustrating the current Iout (at Vout0) versus Vcc to illustrate responses of Iout for different current references. 0013 FIG. 8 is a graph illustrating the current Iout (at Vout0) versus Vcc to illustrate DC and AC responses of Iout for resistive current references. 0014 FIG. 9 is a circuit schematic diagram showing one embodiment of obtaining an offset voltage at the output of the scalable driver of the present invention. 0015 FIG. 10 is a circuit schematic diagram showing another embodiment of obtaining an offset voltage at the output of the scalable driver of the present invention. 0016 FIG. 11 is a waveform diagram illustrating the offset when the circuit of FIG. 9 or 10 is utilized to generate an offset. 0017 FIG. 12 is a circuit schematic diagram illustrating a use of a series resistor in the output line when the load resistor is placed near the scalable driver. 0018 FIG. 13 is a circuit schematic diagram illustrating a use of a series resistor in the output line when the load resistor is placed far from the scalable driver. 0019 FIG. 14 is another embodiment of the present invention in which a transistor network is placed in the mirror leg of the current mirror of the circuit of FIG. 3 in order to provide programmable voltage scaling. 0020 FIG. 15 is an example system level block diagram illustrating the use of the scalable driver to scale various clocking signals from a clock generator.

DETAILED DESCRIPTION OF THE INVENTION 0021 Referring to FIG. 1, a block schematic diagram showing one embodiment of the present invention is illustrated. A current reference or source 10, which provides a current Iout provides a substantially constant driving current to switch 1. The switch 11 is driven by the current source 10 but is switched by an input signal coupled to the switch 11. The current Iout is then coupled through the switch 11 to a load 12 by the switching operation controlled by the input signal (shown as INPUT). The load 12 provides appropriate loading to generate a voltage output (shown as OUTPUT), whenever the current Iout is switched to the load 12 by the operation of the switch 11. In an embodiment described below, the switch 11 is comprised of symmetrical switches. However, it is understood that a variety of switch designs can be implemented for switch 11 to practice the invention. 0022 As noted in FIG. 1, a voltage scaling circuit 13 is used to scale the current so that the output signal developed across the load 12 is less in peak amplitude to the corresponding input signal coupled to the switch 11. The voltage scaling circuit 13 can be a separate circuit coupled to the current source 10 or, alternatively, could be part of the current source 10 itself. Accordingly, when the circuit of FIG. 1 is operational, voltage reduction of the input signal is obtained at the output. The circuit of FIG. 1 operates as a driver and provides voltage scaling, so that one supply voltage platform can drive a circuit operating a second supply voltage platform. For the embodiments described herein the second supply voltage platform (which is the driven platform) is illustrated operating at a lower supply voltage. 0023 For example, utilizing an example circuitry in which a clock signal is coupled to a processor, a 3.3 volt clock signal from a clock generator chip can be coupled to operate with a processor having a processor core supply voltage of around 1.0 volt or below. The voltage scaling reduction is obtained by the voltage developed across the load 12 when current Iout is switched onto the load by the operation of the switch 11. As noted above, the operation of the switch 11 is controlled by the input signal so that the output scaling is achieved for a given input to provide the voltage scaling. 0024 The utilization of a substantially constant current source 10, which is immune to supply noise, allows for a drive of the load 12, which is then also made immune to supply noise. Furthermore, the constant current source 10 provides a high impedance drive at the output, allowing for various loads to be placed at the output without significantly loading the circuit, which loading could affect the signal at the output. Additionally, over-voltage protection at the output (which is the input to the driven circuit) is typically not necessary, since the switch 11 isolate the higher supply voltage of the input from the lower supply voltage driven circuit at the output. 0025 One example embodiment of the driver circuitry of FIG. 1 is shown in FIG. 2. A current reference or source 19 is coupled to a parallel arrangement of P-type transistors 20, 21 (which are also referred to as Mout and Mout, the used to designate a complement signal). The current source 19 is also coupled to a supply voltage Vcc. The output of the transistors 20, 21 are coupled respectively to a pair of output transmission lines 24, 25 to provide the output signals OUT and OUT. The gates of the transistors 20, 21 are coupled to the input signals noted as IN and IN. The two input signals in this instance are complementary signals in which the indicates the complement signal. A pair of load resistors 22, 23 (noted as resistance RL) are coupled to the respective output lines 24, 25. The other end of the resistors 22, 23 are coupled to the supply return Vss, which is shown as ground in the example. 0026 In reference to FIG. 1, transistors 20, 21 correspond to the switch 11 and provide a symmetrical switching operation. The load resistors 22, 23 correspond to the load 12 and provide the loading function at the output. A current source 19 corresponds to the current source 10. The circuit of FIG. 2 has the current source 19 in series with the parallel arrangement of transistors 20, 21 to switch the current Iout to drive the output lines 24 and 25. It is to be noted that Iout is switched onto output line 24 by control of transistor 20 and Iout is switched onto output line 25 by control of transistor 21. In the example, the output on lines 24, 25 is established as a differential output. 0027 The coupling of Iout onto the output lines 24, 25 results in a current flow through the load resistors 22, 23 so that a voltage is developed across the respective resistor 22, 23. Thus, when transistor 20 is biased to draw Iout when IN goes low, Iout flows through resistor 22 developing a voltage equal to IoutRL. A corresponding differential decrease is noted through transistor 21 and resistor 23. Alternatively, when transistor 21 is biased to draw Iout when the input IN goes low, Iout is switched onto line 25 developing a voltage IoutRL across the load 23. 0028 In one embodiment, Iout is set at 11 milliamps while the value of the load resistance RL is set at 50 ohms, so that a voltage of 0.55 volts (V) is developed across each of the load resistance RL. When lines 24, 25 are treated differentially, a differential output swing of 0-0.55 can be developed across the output lines 24, 25. In another embodiment, 0.7V is developed across each RL to generate a 0-0.7 differential output swing. It is to be noted that the output signal can be configured as single-ended or differential. Since the transistors 20, 21 provide isolation between the input and the output, a driving circuit platfonn operating at a first supply voltage can be coupled at the input and a driven circuit platform operating at a second supply voltage can be coupled at the output 24, 25. For the implementation shown, the driven circuit has a lower supply voltage than the driver circuit. 0029 Also, since the current source 19 provides a substantially constant current onto the output lines 24, 25, the circuit is generally immune from variations in the supply voltage. In typical usage, the load resistance RL is equal to the characteristic impedance Z0 of the transmission line 24, 25. Thus, in the example noted above, RL has a value of 50 ohms which is also the characteristic impedance Z0 of the lines 24, 25. 0030 Referring to FIG. 3, a more detailed driver circuit 30 is illustrated in which a 3.3 volt supply (Vcc) platform provides a voltage reduction of at least 3 to 1 to drive a device having a supply voltage platform of around 1.0 volt or below. Without limiting the invention to these voltages, example voltages for the driven circuit can be in the range of 0.5-1.5V for one embodiment. In the circuit 30 of FIG. 3, the switching transistors and the load resistors are equivalent to those shown in FIG. 2. The current source 19 of FIG. 2 is functionally implemented by the current mirror transistors 31, 32. The reference current Iref flows through transistor 31 and the mirroring current Iout flows through the mirror transistor 32. An operational amplifier (op amp) 33 operates as a voltage follower so that the voltage at the minus input of the op amp 33 (also noted as node 34) is impressed onto the plus input of the op amp 33 (also noted as node 35). 0031 The voltage at node 34 is determined by a resistive voltage divider network comprised of resistors 36 and 37, which in the example have the resistance values of R1 and R2, respectively. By proper selection of the voltage division, a bias voltage is established at node 34, as well as node 35. This bias then sets the drive of the current mirror transistors 31, 32. The decoupling capacitor 41 (C1) is used to decouple noise on the bias drive of transistors 31, 32. A second decoupling capacitor (C2) can also be used at node 42. The biasing circuit of op amp 33 provides the voltage scaling circuit 13 noted in FIG. 1. 0032 Accordingly, if the supply voltage of the first circuit is established at a given value, such as 3.3 volts, then some selected voltage value is chosen as a bias voltage to drive transistor 32. In the example circuit 30, resistors 36, 37 provide a voltage divider network (R1, R2) in which the voltage division of resistances R1 and R2, is noted at node 34 and correspondingly also at node 35. It follows then that the current Iref is determined by the voltage at node 35, divided by the resistance of a reference resistor 39 (noted as Rref). The value of this biasing voltage is selected to obtain a desired Iref and correspondingly Iout. 0033 Thus, by establishing corresponding voltages at nodes 34 and 35 and utilizing a particular value for the reference resistor 39, Iref can be selected to have a particular value. Then, due to the current mirror, Iout would follow Iref, in which the proportionate value of Iout to Iref can be scaled utilizing a scaling factor. When current mirror transistors 31, 32 are identical, then it follows that Iout will equal Iref. 0034 In the particular example circuit 30, the reference resistor (Rref) 39 and the load resistors (RL) 22, 23 are shown external to a chip having the circuit 30 (the boundary of the chip is noted by the dotted line). It is appreciated that the load resistors RL and the reference resistor Rref can be implemented on chip, if desired. However, by allowing the RL and Rref to be placed external to the chip, various values of RL and Rref can be selected to provide the appropriate scaling of the output voltage. The additional transistor 40 operates as a dummy transistor. Since one of the switching transistors 20 or 21 is in series with the mirroring transistor 32, the dummy transistor 40 is placed in series with the Iref transistor 31 in order to provide symmetry in both legs of the current mirror circuit 30. 0035 It is also noted that a variety of circuits can be utilized to generate the desired voltage at node 34. The voltage divider network shown in FIG. 3 is just but one example. Other circuitry, including bandgap circuits, can be utilized to provide a reference voltage at node 34. As will be noted below in the description of FIGS. 6 and 7, the reference voltage should be immune to noise and supply voltage variation, so that Iref and Iout remain fairly constant. 0036 In the example of FIG. 3, circuit 30 operates at the supply voltage of 3.3 volts. Accordingly, the circuit 30 would typically operate at the supply voltage of the higher supply voltage platform, such as a 3.3 volt clock generator. The output is coupled to a lower supply voltage platform, such as the approximate 1.1 volt platform of the processor. Thus, the driver circuit 30 need not operate at a lower supply voltage Vcc as the driver device but it does isolate the driven circuit coupled to the output lines 24, 25 so that the 3.3 volts is not impressed onto the output lines 24, 25. Furthermore, the substantially constant current Iout provides a high impedance drive of the output lines 24, 25. It is also appreciated that the reference current Iref need only be derived once to drive a number of mirrored circuits to provide several sets of outputs. 0037 Furthermore, the reference current Iref and the output current Iout can be scaled to provide appropriate voltage scaling. One way to achieve this scaling is to select an appropriate reference resistor Rref. Another technique to change the scaling is to change the physical dimensions of the transistors 31, 32 so that process and/or dimensional differences in these two transistors provide for the current scaling. 0038 FIG. 4 illustrates a waveform at the output. As noted in the waveform diagram, the signal at the output transitions about a mid-point 45 (which is shown by the dotted line). It is appreciated that if the slew rate of the circuit changes, which change is noted in the wave form of FIG. 5, the crossing point will still remain at the mid-point noted by line 46. The slew rate, noted by the slope of the signals when changing states, can change, but the crossover point will still remain at the mid point of the transitions. Thus, circuit 30 can be utilized to provide an output voltage swing which can vary substantially in peak voltage change but without affecting the relative crossover point of the signal. This ensures that signal state changes at the input of the drive circuit are not distorted at the output, even when experiencing changes in the slew rate. 0039 FIG. 6 illustrates a graphic example in which the output current Iout is graphed versus Vout. An ideal current source would generate a flat curve 50 for any output voltage Vout. However, in actual practice, the driver circuit has some amount of effective output impedance associated with changing the voltage on the transistors. Assuming that the transistors 20, 21, 32 of FIG. 3 are operating in saturation, the effective output impedance of the driver is high. The current is substantially constant in this effective range 51, which is shown in FIG. 6. 0040 However, when the transistors 20, 21 leave saturation, then the changes in the output voltage of the driver are seen more directly by transistor 32 and the output impedance drops as shown in the portion just to the right of the effective range 51. When the transistor 32 drops out of saturation, then the changes in the output voltage of the driver are influenced more directly by the transistor 32 and the output impedance drops quickly as noted in the graph of FIG. 6. Thus, the biasing of the driver circuit 30 should be as such to keep Vout in the effective range 51. 0041 Therefore, the design of the current reference will typically take into account the dependency on the supply voltage. If the reference voltage in the current reference of circuit 30 of FIG. 3 is some Vcc independent reference, such as a bandgap circuit, then the current is independent of Vcc. FIG. 7 shows a graph of Iout (when Vout is 0) versus Vcc. If the reference voltage in the current reference is Vcc dependent, such as a resistor divider circuit, then the DC current is directly dependent on the supply voltage. As shown in FIG. 7, a response of a bandgap circuit is relatively flat over the Vcc range of variation. However, the DC response of the resistor circuit is dependent on supply voltage variations. 0042 The Vcc variable current (due to the resistor reference) may potentially be a problem, if supply noise affected Iout to cause jitter in the output signal. However, with high frequency bypassing, higher frequency noise components can be removed. As noted in FIG. 8, in a graph of Iout (when Vout is 0) versus Vcc, the AC response of a resistive reference source is substantially flat or constant, as compared to the DC response. The operative frequency range for noise rejection will be determined mostly by the op amp 33 and the capacitance values of the bypass capacitors, such as capacitors 41, 42. Accordingly, even with resistive current biasing, the reference current can provide a stabilized voltage at the output by providing rejection of AC noise. 0043 Referring to FIG. 9, one circuit for implementing an offset voltage at the output is shown. In order to introduce an offset voltage, an offset noted as Voffset is utilized. Instead of coupling RL to ground, the termination is made to the offset voltage. The actual value of the offset will be determined by the offset voltage. An equivalent offset can also be obtained by using an offset resistance noted by resistor Roffset, which is shown in FIG. 10. An offset resistance Roffset is placed between ground and the junction of the load resistors RL. The current Iout flowing through the offset resistor generates a voltage which is equivalent to the voltage Voffset in FIG. 9. 0044 FIG. 11 illustrates the result of utilizing such an offset. As noted in the waveform diagram of FIG. 11, the output signals OUT and OUT are raised above the reference by the offset voltage imposed by the circuit in FIG. 9 or 10. It is appreciated that other circuits can be implemented to provide the offset voltage. The offset allows for voltage swings at the output having lower differential value, but having the crossover point (noted by dotted line 47) raised above the base threshold 48, which is ground in the example. 0045 FIGS. 12 and 13 also illustrate another embodiment of the present invention to provide a high AC driving impedance at the output. Assuming that the circuit in FIG. 3 is used to drive the output lines 24, 25, the load resistor RL can be placed anywhere along the transmission line. In FIG. 12, RL is placed close to output terminals 60, 61 of the voltage scaling circuit. In FIG. 13, RL is placed closer to the driven circuit and distant from the pads 60 and 61 of the driver circuit. A situation of this nature is encountered when the scaling circuit of the present invention is placed on a die of one chip and is coupled to the lower voltage platform chip across a distance. For example, the transmission line 24, 25 could be lines a circuit board coupling a clock chip to a processor chip. The scaled voltage output from the scaling circuit would then be transmitted to the processor chip over the lines present on the circuit board. 0046 Thus, the load resistor RL could be placed in the vicinity of the scaling circuit or at the processor or anywhere along the transmission line 24, 25. In one embodiment, an external series resistance 62 (also noted as resistor RS) is placed in series with the transmission lines 24, 25 and proximal to the pads 60, 61. In one embodiment where the characteristic impedance and the load resistance RL has a 50 ohm termination, RS is set at 33 ohms. The series resistance RS ensures that the capacitance on the terminal (pin) and silicon of the driver output is not visible to the network and essentially removes the silicon capacitance of the die from the transmission line. This ensures that the termination of the transmission line is achieved with less capacitance, which capacitance may be present due to the length of the transmission line from the wafer die. Thus, where capacitance may play a role in changing the AC impedance at the output of the scaling circuit, the series resistance RS reduces the effect of this capacitance on the transmission line. 0047 FIG. 14 shows another embodiment of the present invention in which programmable scaling is achieved for the scaling circuit shown in FIG. 3. The circuit shown in FIG. 14 is equivalent to the circuit of FIG. 3 except now there is present a scaling circuit 70 which is used to programmably select the value of the current Iout. In the example circuit, transistors 71 (shown as transistors 71a-72c) function as the current mirror transistor equivalent to transistor 32 of FIG. 3. Corresponding transistors 72 (72a-72c) in each leg of the network function as selection transistors, which are selected by the signal S1-S3. 0048 It is appreciated that the selection signal can be coupled to operate in various different modes. For example, in one mode, transistors 72 operate with physical straps to turn on or turn off the transistor in each leg depending on the placement of the strap. In another mode, the selection signals are coupled to a programming source, such as a processor, so that the activation of each leg of the network 70 can be determined by a program. The programming can either set the activation status of each leg of the network 70 when the device is manufactured, at system power on, the processor can change the values on the fly. 0049 With proper scaling of the transistors 71, the scaling between Iref and Iout can be controlled by controlling the number of legs which are activated. It is appreciated that only three legs are shown in the network 70 but the actual number can vary depending on the type of scaling desired. A second dummy transistor 79 is also shown to balance the circuit. Furthermore, it is appreciated that the scaling network can be developed for the reference side of the current mirror. That is, an equivalent network can be put in place for the one current mirror transistor 31. In another embodiment, such networks can be employed at both the reference and the mirror side of the current mirror. 0050 Accordingly, it is appreciated that a variety of techniques can be used to set the scaling between the reference current and the mirrored output current. However, generally, if such programmable network is to be used, it is generally utilized on the mirrored side so that Iref stays substantially constant no matter what the scaling. It is also appreciated that the scaling need not be at integer level and that fractional scaling is also available with the various voltage scaling techniques described herein. 0051 Referring to FIG. 15, a system level diagram is shown in which the scaling driver circuit of the present invention is utilized. In the exemplary system 80 shown in FIG. 15, a clock generator 81 is shown coupled to various devices in which clocking signals are generated and coupled to those various devices. As shown, one clocking signal is coupled to a central processing unit (CPU) 82. The CPU 82, a memory 84 and a graphics controller 87 are coupled to a bridge 83. The bridge 83 is also coupled to a second bridge 85 which couples to various input/output (I/O) devices 86. 0052 As noted, various clock signals are generated by the clock generator 81 and coupled to the various circuits 82, 83, 84, 85, 86, 87. The scaling driver circuit of the present invention, shown by blocks 90a-e are utilized to generate necessary scaled clock signals to the functional components 82-87. It is to be noted that the functional components can be within the clock generator 81 or each individual components can be incorporated in the corresponding target unit. 0053 Using the earlier described example, the clock generator 81 operates on a 3.3 volt supply voltage and the scaling driver circuit shown by circuit 90b scales the 3.3 volts to provide about 1.0 v or below 1.0 v clocking signal to the CPU 82. It is appreciated that the other scaling circuitry 90a, c-e shown in FIG. 15 may not be required if the receiving circuit operates at the same supply voltage level as the clock generator circuit 81. However, FIG. 15 does show that such scaling circuits 90 can be implemented for various devices of a system, such as a computer system. Although the various blocks 90a-e are shown separately, it is appreciated that the circuitry can be implemented as one circuit. As previously described, one reference current could supply a number of mirrored circuits to provide different voltages for the different devices shown. 0054 Additionally, it is to be noted that the scaling circuit need not necessarily be employed with a clocking generator or clocking signals only. For example, if there are data transmissions between circuits operating at two different supply voltage platforms, then the scaling circuit can be utilized to provide the voltage scaling of data transmission. Thus, a data transfer between a CPU and some other component such as bridge 83 of FIG. 15 would be effected over a bus 88. In such circumstance, the scaling circuit could be implemented along bus 88 or at either end of bus 88 to ensure that proper signal scaling is achieved between platforms operating at two different supply voltages. 0055 Various advantages are noted by the practice of the present invention. For example, voltage domain problems are eliminated when a platform component on a higher voltage process needs to drive a receiver, such as a CPU, which is on a lower core voltage platform. The invention allows the driver to drive at an arbitrarily low voltage necessary for the receiver. Low voltage and particularly differential application by the practice of the present invention allows superior EMI performance while lowering jitter and skew. The voltage scalability of the invention allows the platform device to drive a small voltage but still have a high Vcc compared to the driven device. The driven device does not require over voltage protection circuitry which would add timing and performance impact on the receiver. The invention also allows for alternative techniques of programming the scalability as well as utilizing an offset for specific applications. 0056 Furthermore, the circuit provides a high impedance source which allows many termination options at the output without impedance discontinuity, including line length, independent source termination and mid-bus or end-bus driving. Source termination can be effectively implemented with an arbitrarily long line, unlike lower impedance drivers. The high impedance nature also allows drivers to be coupled in the middle of a bus and not affect the impedance of the line. The invention is also very easy to model with a linear model since it operates in a very linear region of the transistor curve, and furthermore, the circuitry for the invention can be implemented in CMOS (complimentary metal oxide semiconductor). Implementation in CMOS allows for inexpensive implementation of voltage scalability. 0057 The circuit can be implemented for both differential applications as well as for single-ended applications. Although the invention can be utilized for various signals, one application utilizes the invention for generating clock signals. The scaling of the clock signals to operate on low supply voltage processors, such as the processors utilized in mobile computers, allows clock vendors to continue to produce inexpensive clock chips of older generation to operate with the more recent newer generation processors on low supply voltage platforms. 0058 Thus, high impedance current mode voltage scalable driver is described.