CROSS REFERENCE TO RELATED PATENTS/PATENT APPLICATIONS

The present U.S. Utility Patent Application claims priority pursuant to 35 U.S.C. Â§ 120, as a continuation, to the following U.S. Utility Patent Application which is hereby incorporated herein by reference in its entirety and made part of the present U.S. Utility Patent Application for all purposes:

1. U.S. Utility application Ser. No. 11/385,632, entitled â€œCurrent-controlled CMOS logic family,â€ filed Mar. 21, 2006 now U.S. Pat. No. 7,215,169, which claims priority pursuant to 35 U.S.C. Â§ 120, as a continuation, to the following U.S. Utility Patent Application which is hereby incorporated herein by reference in its entirety and made part of the present U.S. Utility Patent Application for all purposes:

2. U.S. Utility application Ser. No. 11/114,969, entitled â€œCurrent-controlled CMOS logic family,â€ filed Apr. 26, 2005, now U.S. Pat. No. 7,038,516 B2, issued on May 2, 2006, which claims priority pursuant to 35 U.S.C. Â§ 120, as a continuation, to the following U.S. Utility Patent Application which is hereby incorporated herein by reference in its entirety and made part of the present U.S. Utility Patent Application for all purposes:

3. U.S. Utility application Ser. No. 10/143,087, entitled â€œCurrent-controlled CMOS logic family,â€ filed May 9, 2002, now U.S. Pat. No. 6,900,670 B2, issued on May 31, 2005, which claims priority pursuant to 35 U.S.C. Â§ 120, as a continuation, to the following U.S. Utility Patent Application which is hereby incorporated herein by reference in its entirety and made part of the present U.S. Utility Patent Application for all purposes:

4. U.S. Utility application Ser. No. 09/484,856, entitled â€œCurrent-controlled CMOS logic family,â€ filed Jan. 18, 2002, now U.S. Pat. No. 6,424,194 B1, issued on Jun. 23, 2002, which claims priority pursuant to 35 U.S.C. Â§ 119(e) to the following U.S. Provisional Patent Application which is hereby incorporated herein by reference in its entirety and made part of the present U.S. Utility Patent Application for all purposes:

- a. U.S. Provisional Application Ser. No. 60/141,355, entitled â€œCurrent-controlled CMOS logic family,â€ filed Jun. 28, 1999.

BACKGROUND OF THE INVENTION

The present invention relates in general to integrated circuitry, and in particular to complementary metal-oxide-semiconductor (CMOS) logic and circuits with enhanced speed characteristics.

For a number of reasons CMOS is the logic family of choice in today's VLSI devices. Due to the complementary nature of its operation, CMOS logic consumes near zero static power. CMOS also readily scales with technology. These two features are highly desirable given the drastic growth in demand for low power and portable electronic devices. Further, with the computer aided design (CAD) industry's focus on developing automated design tools for CMOS based technologies, the cost and the development time of CMOS VLSI devices has reduced significantly.

The one drawback of the CMOS logic family, however, remains its limited speed. That is, conventional CMOS logic has not achieved the highest attainable switching speeds made possible by modern sub-micron CMOS technologies. This is due to a number of reasons. Referring to

FIG. 1

, there is shown a conventional CMOS inverter 100â€”the most basic building block of CMOS logic. A p-channel transistor 102 switches between the output and the positive power supply Vcc, and an n-channel transistor 104 switches between the output and the negative power supply (or ground). The switching speed in CMOS logic is inversely proportional to the average on resistance (Ron) of the MOS transistor, and the load capacitance CL on a given node (Ï„=RonÃ—C_L). The on resistance Ron is proportional to the transistor channel length L divided by the power supply voltage (i.e., Ronâˆ L/Vcc), while the load capacitance is given by the gate capacitance of the transistor being driven (i.e., WÃ—LÃ—Cox, where Cox is the gate oxide capacitance), plus the interconnect parasitic capacitance C_int. Therefore, with reduced transistor channel lengths L, the switching speed is generally increased. However, this relationship no longer holds in sub-micron technologies. As the channel length L in CMOS technology shrinks into the sub-micron range, the power supply voltage must be reduced to prevent potential damage to the transistors caused by effects such as oxide breakdown and hot-electrons. The reduction of the power supply voltage prevents the proportional lowering of Ron with the channel length L. Moreover, the load capacitance which in the past was dominated by the capacitances associated with the MOS device, is dominated by the routing or interconnect capacitance (C_int) modern sub 0.5 micron technologies. This means that the load capacitance will not be reduced in proportion with the channel length L. Thus, the RC loading which is the main source of delaying the circuit remains relatively the same as CMOS technology moves in the sub-micron range.

As a result of the speed limitations of conventional CMOS logic, integrated circuit applications in the Giga Hertz frequency range have had to look to alternative technologies such as ultra high speed bipolar circuits and Gallium Arsenide (GaAs). These alternative technologies, however, have drawbacks of their own that have made them more of a specialized field with limited applications as compared to silicon MOSFET that has had widespread use and support by the industry. In particular, compound semiconductors such as GaAs are more susceptible to defects that degrade device performance, and suffer from increased gate leakage current and reduced noise margins. Furthermore, attempts to reliably fabricate a high quality oxide layer using GaAs have not thus far met with success. This has made it difficult to fabricate GaAs FETs, limiting the GaAs technology to junction field-effect transistors (JFETs) or Schottky barrier metal semiconductor field-effect transistors (MESFETs). A major drawback of the bipolar technology, among others, is its higher current dissipation even for circuits that operate at lower frequencies.

It is therefore highly desirable to develop integrated circuit design techniques that are based on conventional silicon CMOS technology, but overcome the speed limitations of CMOS logic.

BRIEF SUMMARY OF THE INVENTION

The present invention is directed to apparatus and methods of operation that are further described in the following Brief Description of the Several Views of the Drawings, the Detailed Description of the Invention, and the claims. Other features and advantages of the present invention will become apparent from the following detailed description of the invention made with reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

FIG. 1

shows a conventional CMOS inverter;

FIG. 2

is an inverter/buffer implemented in C³MOS according to an exemplary embodiment of the present invention;

FIG. 3

shows an exemplary C³MOS level shift buffer according to the present invention;

FIGS. 4A and 4B

show exemplary C³MOS implementations for an AND/NAND gate and an OR/NOR gate, respectively;

FIG. 5

shows an exemplary C³MOS implementation for a 2:1 multiplexer;

FIG. 6

shows an exemplary C³MOS implementation for a two-input exclusive OR/NOR gate;

FIG. 7

is a circuit schematic showing an exemplary C³MOS clocked latch according to the present invention;

FIG. 8

is a circuit schematic for an alternate embodiment for a C³MOS flip-flop according to the present invention;

FIG. 9

shows an exemplary C³MOS implementation for a flip-flop using the C³MOS latch of

FIG. 7

;

FIG. 10

shows a block diagram for a circuit that combines C³MOS and conventional CMOS logic on a single silicon substrate to achieve optimum tradeoff between speed and power consumption;

FIG. 11

shows an exemplary circuit application of the C³MOS/CMOS combined logic wherein C³MOS logic is used to deserialize and serialize the signal stream while CMOS logic is used as the core signal processing logic circuitry;

FIG. 12

is a simplified block diagram of a transceiver system that utilizes the C³MOS/CMOS combined logic according to the present invention to facilitate interconnecting high speed fiber optic communication channels.

DETAILED DESCRIPTION OF THE INVENTION

The present invention provides ultra high-speed logic circuitry implemented in silicon complementary metal-oxide-semiconductor (CMOS) process technology. A distinction is made herein between the terminology â€œCMOS process technologyâ€ and â€œCMOS logic.â€ CMOS process technology as used herein refers generally to a variety of well established CMOS fabrication processes that form a field-effect transistor over a silicon substrate with a gate terminal typically made of polysilicon material disposed on top of an insulating material such as silicon dioxide. CMOS logic, on the other hand, refers to the use of complementary CMOS transistors (n-channel and p-channel) to form various logic gates and more complex logic circuitry, wherein zero static current is dissipated. The present invention uses current-controlled mechanisms to develop a family of very fast current-controlled CMOS (or C³MOSâ„¢) logic that can be fabricated using a variety of conventional CMOS process technologies, but that unlike conventional CMOS logic does dissipate static current. C³MOS logic or current-controlled metal-oxide-semiconductor field-effect transistor (MOSFET) logic are used herein interchangeably.

In a preferred embodiment, the basic building block of this logic family is an NMOS differential pair with resistive loads. Referring to

FIG. 2

, there is shown one embodiment for the basic C³MOS inverter/buffer 200 according to the present invention. Inverter/buffer 200 includes a pair of n-channel MOSFETs 202 and 204 that receive differential logic signals D and D# at their gate terminals, respectively. Resistive loads 206 and 208 connect the drain terminals of MOSFETs 202 and 204, respectively, to the power supply Vcc. Drain terminals of MOSFETs 202 and 204 form the outputs OUT# and OUT of the inverter/buffer, respectively. Resistive loads 206 and 208 may be made up of either p-channel MOSFETs operating in their linear region, or resistors made up of, for example, polysilicon material. In a preferred embodiment, polysilicon resistors are used to implement resistive loads 206 and 208, which maximize the speed of inverter/buffer 200. The source terminals of n-channel MOSFETs 202 and 204 connect together at node 210. A current-source n-channel MOSFET 212 connects node 210 to ground (or negative power supply). A bias voltage VB drives the gate terminal of current-source MOSFET 212 and sets up the amount of current I that flows through inverter/buffer 200. In response to the differential signal at D and D#, one of the two input n-channel MOSFETs 202 and 204 switches on while the other switches off. All of current I, thus flows in one leg of the differential pair pulling the drain terminal (OUT or OUT#) of the on transistor down to logic low, while the drain of the other (off) transistor is pulled up by its resistive load toward logic high. At the OUT output this circuit is a buffer, while at the OUT# output the circuit acts as an inverter.

Significant speed advantages are obtained by this type of current steering logic. Unlike the conventional CMOS inverter of

FIG. 1

, when either one of the input MOSFETs 202 or 204 is switching on, there is no p-channel pull-up transistor that fights the n-channel. Further, circuit 200 requires a relatively small differential signal to switch its transistors. This circuit also exhibits improved noise performance as compared to the CMOS inverter of

FIG. 1

, since in the C³MOS inverter/buffer, transistors do not switch between the power supply and the substrate. Logic circuitry based on current-steering techniques have been known in other technologies such as bipolar, where it is called emitter-coupled logic (ECL), and GaAs where it is called source-coupled FET logic (SCFL). This technique, however, has not been seen in silicon CMOS technology for a number of reasons, among which is the fact that CMOS logic has always been viewed as one that dissipates zero static current. The C³MOS logic as proposed by the present invention, on the other hand, does dissipate static current.

The design of each C³MOS logic cell according to the present invention is optimized based on several considerations including speed, current dissipation, and voltage swing. The speed of the logic gate is determined by the resistive load and the capacitance being driven. As discussed above, the preferred embodiment according to the present invention uses polysilicon resistors to implement the load devices. P-channel MOSFETs can alternatively be used, however, they require special biasing to ensure they remain in linear region. Further, the junction capacitances of the p-channel load MOSFETs introduce undesirable parasitics. Speed requirements place a maximum limit on the value of the resistive loads. On the other hand, the various C³MOS logic cells are designed to preferably maintain a constant voltage swing (IÃ—R). Accordingly, the values for R and I are adjusted based on the capacitive load being driven to strike the optimum trade-off between switching speed and power consumption.

The C³MOS logic family, according to the present invention, contains all the building blocks of other logic families. Examples of such building blocks include inverters, buffers, level shift buffers, N-input NOR and NAND gates, exclusive OR (XOR) gates, flip flops and latches, and the like.

FIG. 3

shows an exemplary C³MOS level shift circuit 300 according to the present invention. Level shift circuit 300 includes essentially the same circuit elements as inverter/buffer 200 shown in

FIG. 2

, with an additional resistor Rs 302 inserted between the power supply Vcc and the load resistors. Circuit 300 operates in the same fashion as inverter/buffer 200 except that it has its power supply voltage shifted by a value equal to (IÃ—Rs). The C³MOS logic circuitry according to the present invention employs this type of level shifter to make the necessary adjustments in the signal level depending on the circuit requirements. Examples of C³MOS circuits utilizing this type of level shifting will be described below in connection with other types of C³MOS logic elements.

FIGS. 4A and 4B

show exemplary C³MOS implementations for an exemplary 2-input AND/NAND gate 400 and an exemplary 2-input OR/NOR gate 402, respectively. These gates operate based on the same current steering principal as discussed above. A logic low signal at input B of AND/NAND gate 400 brings OUT to ground via Q4 while OUT# is pulled high by its load resistor. A logic low at the A input also pulls OUT to ground via Q2 and Q3 (B=high). OUT is pulled high only when both A and B are high disconnecting any path to ground. OUT# provides the inverse of OUT. OR/NOR gate 402 operates similarly to generate OR/NOR logic at its outputs. When another set of transistors are inserted in each leg of the differential pair as is the case for gates 400 and 402, the signals driving the inserted transistors (Q3, Q4) need level shifting to ensure proper switching operation of the circuit. Thus, high speed C³MOS level shifters such as those presented in

FIG. 3

can be employed to drive signals B and B#. In a preferred embodiment, since node OUT in both gates 400 and 402 must drive the additional parasitics associated transistors Q4, dummy load transistors DQL1 and DQL2 connect to node OUT# to match the loading conditions at both outputs.

FIG. 5

shows an exemplary C³MOS implementation for a 2:1 multiplexer 500. Similar to the other C³MOS logic gates, multiplexer 500 includes a differential pair for each input, but multiplexer 500 further includes select transistors 502 and 504 inserted between the common source terminals of the differential pairs and the current source transistor in a cascade structure. By asserting one of the select input signals SELA or SELB, the bias current is steered to the differential pair associated with that select transistor. Thus, signal SELA steers the bias current to the differential pair with A and A# inputs, and signal SELB steers the bias current to the differential pair with B and B# inputs. Similar to gates 400 and 402, the signals SELA and SELB driving 15 inserted transistors 502 and 504 need level shifting to ensure proper switching operation of the circuit.

FIG. 6

shows an exemplary C³MOS implementation for a two-input exclusive OR (XOR) gate 600. This implementation includes two differential pairs 602 and 606 that share the same resistive load, receive differential signals A and A# at their inputs as shown, and have their drain terminals cross-coupled at the outputs. The other differential input signals B and B# are first level shifted by circuit 606 and then applied to cascade transistors 608 and 610 that are inserted between the differential pairs and the current source transistor. The circuit as thus constructed performs the XOR function on the two input signals A and B.

FIG. 7

is a circuit schematic showing an exemplary C³MOS clocked latch 700 according to the present invention. Latch 700 includes a first differential pair 702 that receives differential inputs D and D# at the gate terminals, and a second differential pair 704 that has its gate and drain terminals cross-coupled to the outputs of OUT and OUT# first differential pair 702. Clocked transistors 706 and 708 respectively connect common-source nodes of differential pairs 702 and 704 to the current-source transistor. Complementary clock signals CK and CKB drive the gate terminals of clocked transistors 706 and 708. Similar to the other C³MOS gates that have additional transistors inserted between the differential pair and the current-source transistor, clock signals CK and CKB are level shifted by level shift circuits such as that of

FIG. 3

.

A C³MOS master-slave flip-flop 800 according to the present invention can be made by combining two latches 700 as shown in

FIG. 8

. A first latch 802 receives differential input signals D and D# and generates differential output signals QI and QI#. The differential output signals QI and QI# are then applied to the differential inputs of a second latch 804. The differential outputs Q and Q# of second latch 804 provide the outputs of flip-flop 800.

Every one of the logic gates described thus far may be implemented using p channel transistors. The use of p-channel transistors provides for various alternative embodiments for C³MOS logic gates.

FIG. 9

shows one example of an alternative implementation for a C³MOS clocked latch 900 that uses p-channel transistors. In this embodiment, instead of inserting the n-channel clocked transistors between the common-source nodes of the differential pairs and the current-source transistor, p channel clocked transistors 902 and 904 connect between the common-source nodes and the power supply Vcc. This implementation also requires that each differential pair have a separate current-source transistor as shown. Clocked latch 900 operates essentially the same as latch 700 shown in

FIG. 7

, except the implementation is not as efficient both in terms of size and speed.

As illustrated by the various C³MOS logic elements described above, all of the building blocks of any logic circuitry can be constructed using the C³MOS technique of the present invention. More complex logic circuits such as shift registers, counters, frequency dividers, etc., can be constructed in C³MOS using the basic elements described above. As mentioned above, however, C³MOS logic does consume static power. The static current dissipation of C³MOS may become a limiting factor in certain large scale circuit applications. In one embodiment, the present invention combines C³MOS logic with conventional CMOS logic to achieve an optimum balance between speed and power consumption. According to this embodiment of the present invention, an integrated circuit utilizes C³MOS logic for the ultra high speed (e.g., GHz) portions of the circuitry, and conventional CMOS logic for the relatively lower speed sections. For example, to enable an integrated circuit to be used in ultra high speed applications, the input and output circuitry that interfaces with and processes the high speed signals is implemented using C³MOS. The circuit also employs C³MOS to divide down the frequency of the signals being processed to a low enough frequency where conventional CMOS logic can be used. The core of the circuit, according to this embodiment, is therefore implemented by conventional CMOS logic that consumes zero static current.

FIG. 10

shows a simplified block diagram illustrating this exemplary embodiment of the invention. A C³MOS input circuit 1000 receives a high frequency input signal IN and outputs a divided down version of the signal IN/n. The lower frequency signal IN/n is then processes by core circuitry 1002 that is implemented in conventional CMOS logic. A C³MOS output circuit 1004 then converts the processed IN/n signal back to the original frequency (or any other desired frequency) before driving it onto the output node OUT.

An example of a circuit implemented using combined CMOS/C³MOS logic according to the present invention is shown in

FIG. 11

. C³MOS input circuitry 1100 is a deserializer that receives a serial bit stream at a high frequency of, for example, 2 GHz. A 2 GHz input clock signal CLK is divided down to 1 GHz using a C³MOS flip-flop 1102, such as the one shown in

FIG. 8

, that is connected in a Ã·2 feedback configuration. The 1 GHz output of flip-flop 1102 is then supplied to clock inputs of a pair of C³MOS latches 1104 and 1106. Latches 1104 and 1106, which may be of the type shown in

FIG. 6

, receive the 2 GHz input bit stream at their inputs and respectively sample the rising and falling edges of the input bit stream in response to the 1 GHz clock signal CLKI2. The signal CLKI2 which is applied to the B/B# inputs of each latch (the level shifted input; see

FIG. 6

), samples the input data preferably at its center. It is to be noted that the rise and fall times of the signal in CMOS logic is often very dependent on process variations and device matching. C³MOS logic, on the other hand, is differential in nature and therefore provides much improved margins for sampling.

Referring back to

FIG. 11

, block 11 thus deserializes the input bit stream with its frequency halved to allow for the use of conventional CMOS logic to process the signals. The signals at the outputs of latches 1104 and 1106 are applied to parallel processing circuitry 1108 that are implemented in conventional CMOS logic operating at 1 GHz. The reverse is performed at the output where a serializer 1110 receives the output signals from processing circuitry 1108 and serializes them using C³MOS logic. The final output signal is a bit stream with the original 2 GHz frequency. Circuit applications wherein this technique can be advantageously employed include high speed single or multi-channel serial links in communication systems.

As apparent from the circuit shown in

FIG. 11

, this technique doubles the amount of the core signal processing circuitry. However, since this part of the circuit is implemented in conventional CMOS logic, current dissipation is not increased by the doubling of the circuitry. Those skilled in the art appreciate that there can be more than one level of deserializing if further reduction in operating frequency is desired. That is, the frequency of the input signal can be divided down further by 4 or 8 or more if desired. As each resulting bit stream will require its own signal processing circuitry, the amount and size of the overall circuitry increases in direct proportion to the number by which the input signal frequency is divided. For each application, therefore, there is an optimum number depending on the speed, power and area requirements.

According to one embodiment of the present invention the combined C³MOS/CMOS circuit technique as shown in

FIG. 11

is employed in a transceiver of the type illustrated in

FIG. 12

. The exemplary transceiver of

FIG. 12

is typically found along fiber optic channels in high speed telecommunication networks. The transceiver includes at its input a photo detect and driver circuit 1200 that receives the input signal from the fiber optic channel. Circuit 1200 converts fiber-optic signal to packets of data and supplies it to a clock data recovery (CDR) circuit 1202. CDR circuit 1202 recovers the clock and data signals that may be in the frequency range of about 2.5 GHz, or higher. Established telecommunication standards require the transceiver to perform various functions, including data monitoring and error correction. These functions are performed at a lower frequency. Thus, the transceiver uses a demultiplexer 1204 which deserializes the 2.5 GHz data stream into, for example, 16 parallel signals having a frequency of about 155 MHz. An application specific integrated circuit (ASIC) 1206 then performs the monitoring and error correction functions at the lower (155 MHz) frequency. A multiplexer and clock multiplication unit (CMU) 1208 converts the parallel signals back into a single bit stream at 2.5 GHz. This signal is then retransmitted back onto the fiber optic channel by a laser drive 1212. The combined C³MOS/CMOS technique of the present invention allows fabrication of demultiplexer 1204, ASIC 1206 and multiplexer and CMU 1208 on a single silicon die, as indicated by reference numeral 1210, in a similar fashion as described in connection with the circuit of

FIGS. 10 and 11

. That is, demultiplexer 1204 and multiplexer and CMU 1208 are implemented in C³MOS with ASIC 1206 implemented in conventional CMOS.

In conclusion, the present invention provides various circuit techniques for implementing ultra high speed circuits using current-controlled CMOS (C³MOS) logic fabricated in conventional CMOS process technology. An entire family of logic elements including inverter/buffers, level shifters, NAND, NOR, XOR gates, latches, flip-flops and the like have been developed using C³MOS according to the present invention. In one embodiment, the present invention advantageously combines high speed C³MOS logic with low power conventional CMOS logic. According to this embodiment circuits such as transceivers along fiber optic channels can be fabricated on a single chip where the ultra-high speed portions of the circuit utilize C³MOS and the relatively lower speed parts of the circuit use conventional CMOS logic. In another embodiment, the C³MOS logic circuitry receives a first power supply voltage that is higher than the power supply voltage used by the conventional CMOS logic circuitry. While the above is a complete description of the preferred embodiment of the present invention, it is possible to use various alternatives, modifications and equivalents. Therefore, the scope of the present invention should be determined not with reference to the above description but should, instead, be determined with reference to the appended claims, along with their full scope of equivalents.